Electric-power conversion apparatus

ABSTRACT

There is provided an electric-power conversion apparatus including a chopper circuit; a current sense resistor that detects the output current of the chopper circuit; a differential detection circuit that outputs, as a differential detection signal (vo), the electric potential difference across the current sense resistor; and a calculation means that corrects the differential detection signal (vo) from the differential detection circuit by use of a control signal (D 1 ) for the chopper circuit so as to calculate the output current (i 0 ) of the chopper circuit.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to an electric-power conversion apparatusprovided with an electric current detection function capable ofaccurately calculating the value of the output current of a choppercircuit for supplying electric power to a load.

2. Description of the Related Art

Most of electric-power conversion apparatuses such as inverters orconverters each include, as a chopper circuit, an upper stage powersemiconductor device connected between a first electric potential and asecond electric potential and a lower stage power semiconductor deviceconnected between the second electric potential and a reference electricpotential. Based on a desired phase width signal (duty ratio), at leastone of the upper stage power semiconductor device and the lower stagepower semiconductor device is turned on and off so that electric-powerconversion is performed between the first electric potential and thesecond electric potential. For example, in the case where this kind ofelectric-power conversion apparatus is applied to an inverter thatdrives a three-phase AC rotating machine, a DC power source is connectedbetween the first electric potential and the reference electricpotential, a desired phase width is appropriately given, and based onthe phase width, the upper stage power semiconductor device and thelower stage power semiconductor device are switched, so that the secondelectric potential with respect to the reference electric potential isconverted into an AC voltage. In other words, electric-power conversionis performed in such a way that the first electric potential, which is aDC voltage with respect to the reference electric potential, isconverted into the second electric potential, which is an AC voltage. Inan inverter that drives a three-phase AC rotating machine, by applyingthis kind of electric-power conversion to each of the U-phase, theV-phase, and the W-phase, electric-power conversion from a DC power intoa three-phase AC power is performed.

In the case where a three-phase AC rotating machine is driven, anelectric current supplied by the electric-power conversion apparatus andthe torque of the three-phase AC rotating machine are in a closerelationship; therefore, insufficient accuracy of detecting the electriccurrent causes a torque ripple, rotational speed ripple, or abnormalnoise in the three-phase AC rotating machine.

For example, in the case where there is driven a three-phase AC rotatingmachine provided in an electric power steering apparatus mounted in avehicle, a torque ripple emerges as a vibration of the steering wheel;therefore, it is required to accurately detect the electric current soas to reduce the torque ripple as much as possible.

The methods of detecting the output current of an electric-powerconversion apparatus include a method utilizing a current transformerand a method utilizing a hole current sensor. Because being not capableof detecting a DC current, the method utilizing a current transformercannot be adopted in the application in which a three-phase AC rotatingmachine is in the stop mode or driven at a low rotation speed. Inaddition, when being mounted in a vehicle, an electric-power conversionapparatus needs to operate at a severe temperature, at a severehumidity, in a vibration, and in a dusty condition; thus, a currentdetecting resistor is superior to a hole device as the hole currentsensor in terms of robustness.

Accordingly, in a conventional electric-power conversion apparatusdisclosed in Patent Document 1, there are provided arm circuits forthree phases in each of which an upper-arm switching device and alower-arm switching device are connected in series with each other, theupper-arm switching device is connected with the positive electrode of aDC power source, and the lower-arm switching device is connected withthe negative electrode of the DC power source; and current detectionresistors that are inserted in series into the respective lower-armswitching devices for at least two phases and each of which detects acurrent flowing in the lower-arm switching device. By comparing acarrier and a three-phase voltage command wave signal generated based onthe foregoing current, the upper-arm and lower-arm switching devices areon/off-controlled through pulse-width modulation, so that a current tobe supplied by electric-power conversion apparatus is obtained.

Moreover, for example, a conventional electric-power conversionapparatus disclosed in Patent Document 2 is provided with a currentdetection means having a shunt resistor provided at the power sourceside of a load to be PWM-controlled by a drive circuit; in each of theon duration and the off duration of a PWM control signal, the voltageacross the shunt resistor is detected; based on the difference betweenthe voltage detected during the on duration and an offset voltage, whichis the voltage detected during the off duration, the current flowing ina load is detected.

Furthermore, for example, in a conventional electric-power conversionapparatus disclosed in Patent Document 3, a current current senseresistor is inserted into the output circuit of an inverter that invertsDC electric power into AC electric power for driving a motor, by use ofa power device having a gate circuit, and there are provided asigma-delta modulator that converts the voltage generated across thecurrent sense resistor into a sigma-delta modulated digital signal, anelectricity insulating coupler that transfers the digital signal in anelectrically insulated manner, a digital filter that demodulates thetransferred signal, and a control signal generation circuit thatcontrols the gate circuit of the inverter, based on the output of thedigital filter, so that the current to be supplied by the electric-powerconversion apparatus is obtained.

Still moreover, for example, in a conventional electric-power conversionapparatus disclosed in Patent Document 4, a motor is driven by anupper-arm switching device and a lower-arm switching device, and thereare provided a first reference voltage source that adopts, as areference, the electric potential of the positive electrode of thebattery for the motor, a second reference voltage source that adopts, asa reference, the electric potential of the ground, a shunt resistorprovided between a terminal of the motor and the connection pointbetween the upper-arm switching device and the lower-arm switchingdevice, a switching means that outputs the voltage from one of the firstand second reference voltage sources, based on the electric potential ofthe shunt resistor, a voltage dividing means that divides the output ofthe switching means and the voltage across the shunt resistor so as togenerate two voltage signals, and an amplification means that receivesthe two voltage signals generated by the voltage dividing means andoutputs a voltage that increases or decreases at a predeterminedamplification factor in accordance with the value and the direction of acurrent flowing in the shunt resistor, so that the current to besupplied by the electric-power conversion apparatus is obtained.

PRIOR ART REFERENCE Patent Document

-   [Patent Document 1] Japanese Patent Application Laid-Open No.    2009-017671-   [Patent Document 2] Japanese Patent Application Laid-Open No.    2006-262677-   [Patent Document 3] Japanese Patent Application Laid-Open No.    H7-015972-   [Patent Document 4] Japanese Patent Application Laid-Open No.    2006-064596

In these conventional electric-power conversion apparatuses, there havebeen the following problems.

In the conventional electric-power conversion apparatus disclosed inPatent Document 1, because the current detection resistors that detectcurrents are inserted in series into the respective lower-arm switchingdevices, currents flowing in the respective lower-arm switching devicescan be obtained; however, currents flowing in the upper-arm switchingdevices cannot be obtained. Accordingly, the current during the durationin which the upper-arm switching device is on and the lower-armswitching device is off cannot be obtained; thus, there has been aproblem that the on/off command for the switching device is restricted.There has been a problem that when a three-phase AC rotating machine isdriven by utilizing this kind of electric-power conversion apparatus,the restriction on the on/off command for the switching device alsorestricts the maximum voltage that can be outputted by theelectric-power conversion apparatus.

Moreover, in the conventional electric-power conversion apparatusdisclosed in Patent Document 2, because the current flowing in the loadis detected by adopting, as an offset voltage for a voltage detectedduring a duration in which the PWM control signal is on, a voltagedetected during a duration in which the PWM control signal is off, thereexists restriction that the shunt resistor needs to be located in a pathin which no current flows while the PWM control signal is off; thus,there has been a problem that there cannot be detected a current flowingin a path in which the current flows even while the PWM control signalis off.

Still moreover, in the conventional electric-power conversion apparatusdisclosed in Patent Document 3, because a current sense resistor isinserted into the output circuit of an inverter that inverts DC electricpower into AC electric power for driving a motor, by use of a powerdevice having a gate circuit, and a current is obtained based on thevoltage across the current detection resistor; therefore, theconventional electric-power conversion apparatus disclosed in PatentDocument 3 has an advantage in that unlike the electric-power conversionapparatus disclosed in Patent Document 1, no restriction is imposed onthe on/off command for the switching device. However, because a digitalsignal is transferred in an electrically insulated manner, there hasbeen a problem that a power source is required also at the insulatedside or that the apparatus is expensive because an electricityinsulating coupler such as a photocoupler is required.

Furthermore, in the conventional electric-power conversion apparatusdisclosed in Patent Document 4, because a current is obtained bydetecting, in an insulating manner, the electric potential differencebetween the shunt resistor provided between both terminals of a motorand the connection point between the upper-arm switching device and thelower-arm switching device, there exists no restriction on the on/offcommand for the switching device, and no power source for the insulationis required. However, in order to configure the switching means, thereare required a PNP-type transistor that outputs a voltage of a firstreference voltage source when the voltage across the shunt resistor isrelatively high and an NPN-type transistor that outputs a voltage of asecond reference voltage source when the voltage across the shuntresistor is relatively low; thus, there has been a problem that theapparatus is expensive.

SUMMARY OF THE INVENTION

The present invention has been implemented in order to solve theforegoing problems; the objective thereof is to obtain an electric-powerconversion apparatus that is inexpensive and is provided with anelectric current detection function capable of accurately calculatingthe output current of a chopper circuit.

An electric-power conversion apparatus according to the presentinvention includes a chopper circuit; a current sense resistor thatdetects the output current of the chopper circuit; a differentialdetection circuit that outputs, as a differential detection signal, theelectric potential difference across the detection resistor; and acalculation means that corrects the differential detection signal fromthe differential detection circuit by use of a control signal for thechopper circuit so as to calculate the output current of the choppercircuit.

The calculation means calculates a switching signal for turning on oroff the chopper circuit based on a phase width signal, which is acontrol signal for the chopper circuit, and calculates the outputcurrent of the chopper circuit based on the differential detectionsignal; the chopper circuit is configured with an upper stage powersemiconductor device connected between a first electric potential and asecond electric potential and a lower stage power semiconductor deviceconnected between the second electric potential and a reference electricpotential; the current sense resistor is connected between the secondelectric potential and the third electric potential; and thedifferential detection circuit detects the electric potential differencebetween the second electric potential and the third electric potential.

In the electric-power conversion apparatus according to the presentinvention, the differential detection signal is corrected by the controlsignal for the chopper circuit; therefore, because there can beperformed calculation of the output current, of the chopper circuit,which corresponds to an offset change caused by an in-phase voltage,there can be obtained an electric-power conversion apparatus that isinexpensive and can accurately detect a current.

In an electric-power conversion apparatus according to the presentinvention, because the current sense resistor is connected between thesecond electric potential and the third electric potential, the currentsupplied from the second electric potential to the third electricpotential can be obtained, regardless of whether or not the upper stagepower semiconductor device or the lower stage power semiconductor deviceis on.

The foregoing and other object, features, aspects, and advantages of thepresent invention will become more apparent from the following detaileddescription of the present invention when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 1 of thepresent invention;

FIG. 2 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 1;

FIG. 3 is a set of charts for explaining the relationship, in thecalculation means according to Embodiment 1, among the current i0 at atime when based on a desired phase width signal D1 and a carrier-wavesignal C1, an upper stage power semiconductor device and a lower stagepower semiconductor device are operated, the differential detectionsignal vo, the differential detection signal error voe, and the signalvof that is obtained by applying lowpass-filter processing to thedifferential detection signal;

FIG. 4 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 2 of thepresent invention;

FIG. 5 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 2;

FIG. 6 is a set of charts for explaining the calculation, by thecalculation means according to Embodiment 2, of the switching signals G1and G2 based on a desired phase width signal D1 and a carrier-wavesignal C1;

FIG. 7 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 3 of the present invention;

FIG. 8 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 4 of the present invention;

FIG. 9 is a set of charts representing the relationship among the timeand the respective signals in the calculation means according toEmbodiment 4;

FIG. 10 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 5 of the present invention;

FIG. 11 is a flowchart representing the operation of a selectoraccording to Embodiment 5;

FIG. 12 is a set of charts representing the relationship among the timeand the respective signals in Embodiment 5;

FIG. 13 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 6 of the present invention;

FIG. 14 is a flowchart representing the operation of a selectoraccording to Embodiment 6;

FIG. 15 is a set of charts representing the relationship among the timeand the respective signals in Embodiment 6;

FIG. 16 is a diagram illustrating the internal configuration of acalculation means according to Embodiment 7 of the present invention;

FIG. 17 is a diagram illustrating the internal configuration of adifferential detection circuit according to Embodiment 8 of the presentinvention;

FIG. 18 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 10 of thepresent invention;

FIG. 19 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 11 of thepresent invention; and

FIG. 20 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 12 of thepresent invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment 1

FIG. 1 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 1 of thepresent invention. In FIG. 1, an electric load 5 and a DC voltage source1 having a voltage difference v1 are connected with an electric-powerconversion apparatus 15. The electric-power conversion apparatus 15 isprovided with a chopper circuit 16, a differential detection circuit 6,a calculation means 13, and a phase width signal generator 14; as thecontrol signals for the chopper circuit 16, there exist a phase widthsignal D1 outputted by the phase width signal generator 14 and aswitching signal S1 outputted by the calculation means 13. The phasewidth signal D1 corresponds to the ratio of the conduction duration ofthe upper stage power semiconductor device 2 to the conduction durationof the lower stage power semiconductor device 3 and varies ranging from“0” to “1”; the upper stage power semiconductor device 2 and the lowerstage power semiconductor device 3 are included in the chopper circuit16. The phase width signal D1 is also referred to as a duty ratio. Theswitching signal S1 is a signal for turning on or off the upper stagepower semiconductor device 2 included in the chopper circuit 16.

The negative electrode of the DC voltage source 1 having the voltagedifference v1 is connected with the ground potential; The positiveelectrode of the DC voltage source 1 is connected with one of theterminals of the upper stage power semiconductor device 2 included inthe chopper circuit 16. The upper stage power semiconductor device 2turns on and off when the switching signal S1 is “TRUE” and “FALSE”,respectively. One of the terminals of the lower stage powersemiconductor device 3 and one of the terminals of a current senseresistor 4 having a resistance value of R0 [Ω] are connected with theother terminal of the upper stage power semiconductor device 2; theother terminal of the lower stage power semiconductor device 3 isconnected with the ground potential. The lower stage power semiconductordevice 3 turns on when a current flows from the ground potential to thelower stage power semiconductor device 3; the lower stage powersemiconductor device 3 turns off when a current flows from the lowerstage power semiconductor device 3 to the ground potential. InEmbodiment 1, the upper stage power semiconductor device 2 is formed ofa semiconductor switching device such as a MOS-FET or an IGBT; the lowerstage power semiconductor device 3 is formed of a semiconductorrectifier such as a diode.

A second electric potential v2 becomes equal to the first electricpotential v1 during a duration in which the upper stage powersemiconductor device 2 is on. The electric load 5 is a load consistingof a coil and a resistor that are connected in series with each other. Acurrent that passes through a coil always varies continuously;therefore, when the state of the upper stage power semiconductor device2 changes from “on” to “off”, the current i0 is supplied from thereference electric potential to the electric load 5 by way of the powersemiconductor device 3, which is a diode. The one terminal of theelectric load 5 is connected with the other terminal of the currentsense resistor 4, and the other terminal of the electric load 5 isconnected with the ground potential.

Here, definition is made in such a way that the ground potential is thereference electric potential, the positive-electrode potential of the DCvoltage source 1 is the first electric potential v1, the electricpotential of the connection point between the upper stage powersemiconductor device 2 and the lower stage power semiconductor device 3is the second electric potential v2, the electric potential of theconnection point between the current sense resistor 4 and the electricload 5 is a third electric potential v3. Additionally, definition ismade in such a way that the current supplied to the electric load 5 byway of the current sense resistor 4 is a current i0. In Embodiment 1,the electric load 5 is a load consisting of a coil and a resistor thatare connected in series with each other; however, there is norestriction on the combination for the electric load; a capacitor, abattery, and the like may be connected in series with and/or in parallelwith one another.

The differential detection circuit 6 detects the electric potentialdifference between the second electric potential v2 and the thirdelectric potential v3 and then outputs a differential detection signalvo. The differential detection circuit 6 is configured with anoperational amplifier 7 and resistors 8 through 12. The resistor 8having a resistance value of R1 [Ω] is connected between the negativedifferential input terminal of the operational amplifier 7 and the thirdelectric potential v3; the resistor 9 having a resistance value of R2[Ω] is connected between the negative differential input terminal of theoperational amplifier 7 and the output terminal of the operationalamplifier 7. The resistor 10 having a resistance value of R3 [Ω] isconnected between the positive differential input terminal of theoperational amplifier 7 and the second electric potential v2; theresistor 11 having a resistance value of R4 [Ω] is connected between thepositive differential input terminal of the operational amplifier 7 andan arbitrary electric potential vcc; the resistor 12 having a resistancevalue of R5 [Ω] is connected between the positive differential inputterminal of the operational amplifier 7 and the ground potential. Theresistors 8 through 12 are selected in such a way that the followingequations (1) and (2) are satisfied among the resistance values R1through R5.

R1=R3  (1)

2×R2=R4=R5  (2)

The arbitrary electric potential vcc determines the offset value of thedifferential detection signal vo, which is the output of the operationalamplifier 7; the differential detection signal vo varies with respect tovcc/2, in accordance with the current i0. In the case where the currenti0 is “0”, the output of the differential detection signal vo becomesvcc/2 [V]. For example, in the case where vcc is set to the foregoingreference electric potential, i.e., the ground potential, thedifferential detection signal vo varies with respect to the referenceelectric potential, ranging from a positive value to a negative value inaccordance with the sign of the current i0. In the case where vcc is setto 5 [V], the differential detection signal vo varies with respect to2.5 [V], in accordance with the current i0. Furthermore, when beingconfigured to be non-insulated from the ground potential, thedifferential detection circuit 6 does not require both positive andnegative power sources; thus, as the power source for the differentialdetection circuit 6, only an inexpensive positive power source isrequired.

The calculation means 13 calculates a switching signal S1 for turning onor off the upper stage power semiconductor device 2, based on a desiredphase width signal D1 obtained from the phase width signal generator 14,calculates the current i0 passing through the current sense resistor 4,based on the differential detection signal vo, and outputs the currenti0 as a calculation current i1.

FIG. 2 is a diagram illustrating the internal configuration of thecalculation means 13 according to Embodiment 1. In FIG. 2, acarrier-wave generator 20 outputs a triangular-shaped carrier-wavesignal C1 having the minimum value of “0”, the maximum value “1”, andthe cycle of Tc [sec]. The cycle Tc [sec] is also referred to as thecarrier period in a PWM modulation. A comparator 21 outputs the signalS1 having a value of “TRUE” or “FALSE”. The comparator 21 compares thelevel of the phase width signal D1 and the level of the carrier-wavesignal C1; in the case where D1 C1, the comparator 21 outputs “TRUE” asthe signal S1, and in the case where D1<C1, the comparator 21 outputs“FALSE” as the signal S1.

A gain calculator 25 multiplies the phase width signal D1 by a gain Koffand then outputs the result of the multiplication as a corrected signalvoc.

A lowpass filter 26 outputs, as the signal vof, a signal obtained byapplying lowpass-filter processing to the differential detection signalvo. The time constant of the lowpass filter 26 is set to a constantraging from a value that is approximately equal to the cycle Tc of thecarrier-wave signal C1 to a value that is approximately 20 times aslarge as the cycle Tc. The setting of the time constant will beexplained later.

An adder-subtractor 27 adds an offset signal voff obtained from anoffset setter 28 to the signal vof obtained through the lowpass-filterprocessing and subtracts the corrected signal voc obtained from the gaincalculator 25 from the signal vof; then, the adder-subtractor 27 outputsthe result of the addition and the subtraction to a gain calculator 29.The gain calculator 29 multiplies the output of the adder-subtractor 27by “K” and outputs the result of the multiplication, as the calculationcurrent i1.

FIG. 3 is a set of charts for explaining the relationship, in thecalculation means 13, among the current i0 at a time when based on thedesired phase width signal D1 and the carrier-wave signal C1, the upperstage power semiconductor device 2 is operated, the differentialdetection signal vo, the differential detection signal error voe, andthe signal vof that is obtained by applying lowpass-filter processing tothe differential detection signal vo.

In FIG. 3, there is dealt with a case where as the time elapses, thedesired phase width signal D1 monotonically increases; however, when thedesired phase width signal D1 is set to an AC signal alternating withrespect to “0.5”, the second electric potential may be an AC voltagealternating with respect to the middle potential between the firstelectric potential and the reference voltage. In FIG. 3, the top chartrepresents the waveforms of the phase width signal D1 and thecarrier-wave signal C1. The phase width signal D1 ranges from “0” to“1”. The carrier-wave signal C1 has the minimum value “0”, the maximumvalue “1”, and the cycle Tc. The second top chart represents thewaveform of the switching signal S1. The comparator 21 compares thelevel of the phase width signal D1 and the level of the carrier-wavesignal C1; in the case where D1 C1, the signal S1 becomes “TRUE”, and inthe case where D1<C1, the signal S1 becomes “FALSE”.

The second electric potential v2 approximately coincides with a valueobtained by multiplying the signal S1 by the first electric potentialv1. As a result, the current i0 has such a waveform as represented inthe third top chart in FIG. 3. The differential detection signal vovaries with respect to vcc/2 [V], in accordance with the current i0, andhas such a waveform as represented in the fourth top chart in FIG. 3.The waveform of the differential detection signal vo in the fourth topchart in FIG. 3 is the sum of the value that is proportional to thecurrent i0 and the value that is proportional to the switching signalS1; the center of the waveform is vcc/2 [V]. When the differentialdetection circuit 6 operates ideally, the differential detection signalvo has a waveform with the center of vcc/2 [V], in proportion to thecurrent i0; however, because in practice, complete common mode rejectioncannot be implemented, the differential detection signal vo has such awaveform as represented in the fourth top chart in FIG. 3.

Here, the common mode rejection will be explained. The equation (3)below is established among the current i0, the second electric potentialv2, the arbitrary electric potential vcc, and the differential detectionsignal vo.

vo=K1×i0+K2×v2+K3×vcc  (3)

where K1, K2, and K3 are constants that are defined by the equations (4)through (6) below.

K1=−(R2÷R1)×R0  (4)

K2={R1×R4×R5−R2×R3×(R5+R4)}÷{R1×(R3×R4+R3×R5+R4×R5)}  (5)

K3={(R1+R2)×R3×R5}÷{R1×(R3×R4+R3×R5+R4×R5)}  (6)

The gain K utilized for multiplication by the gain calculator 29corresponds to 1/K1 in the equation (3). Accordingly, K may be expressedby “−R1÷(R2×R0)”.

In particular, in the case where the relationships expressed by theequations (1) and (2) are established among R1 through R5, K2=0 andK3=½; thus, the differential detection signal vo is given by theequation (7) below.

vo=K1×i0+vcc/2  (7)

However, when the differential detection circuit 6 is actuallyconfigured, there exist unevennesses in the resistance values R1 throughR5. As a result, because in the equation (5), K2≠0, the differentialdetection signal vo is given by the equation (3). If the second item inthe right-hand side of the equation (3) can be removed, the differentialdetection signal vo has a desirable waveform that is proportional to thecurrent i0 and whose center line is vcc/2. Removal of the second item inthe right-hand side of the equation (3) is the common mode rejection.

The fifth top chart in FIG. 3 represents the differential detectionsignal error voe; in the case where K1×i0 v2, the second item in theright-hand side of the equation (3) becomes dominant, and hence theequation (8) below is established.

voe K2×v2  (8)

The bottom chart in FIG. 3 represents the signal vof obtained byapplying lowpass-filter processing to the differential detection signalvo. The signal vof is the same signal as obtained by applyinglowpass-filter processing to the right-hand side of the equation (3).Because K1 is a constant value, when undergoing lowpass-filterprocessing, the first item in the right-hand side of the equation (3)becomes a value obtained by multiplying the filter-processed current i0by “K1”. Because K2 is also a constant value, when undergoinglowpass-filter processing, the second item in the right-hand side of theequation (3) becomes a value obtained by multiplying thefilter-processed differential detection signal error voe by “K2”.

Here, there will be considered the value obtained by applyinglowpass-filter processing to the differential detection signal errorvoe. The differential detection signal error voe is proportional to thesecond electric potential v2. The second electric potential v2 iscontrolled by the chopper circuit 16 in such a way that the averagevalue thereof coincides with a value obtained by multiplying the phasewidth signal D1 by the first electric potential v1. The time constant ofthe lowpass filter 26 in Embodiment 1 is set to a constant raging from avalue that is approximately equal to the cycle Tc of the carrier-wavesignal C1 to a value that is approximately 20 times as large as thecycle Tc. By applying a filter having a time constant that isapproximately equal to the cycle Tc or larger to the differentialdetection signal error voe, the differential detection signal error voe,which is proportional to the second electric potential v2, is alsoaveraged and becomes a value proportional to the phase width signal D1.

If the time constant of the lowpass filter 26 is approximately 20 timesas large as the cycle Tc, it is large enough to average the differentialdetection signal error voe; if the time constant becomes larger thanthat, a disadvantage is provided due to delayed detection, althoughthere is demonstrated an advantage that the differential detectionsignal error voe can be averaged. As described above, the value obtainedby applying lowpass-filter processing to the differential detectionsignal error voe can be regarded as being proportional to the phasewidth signal D1.

Because K3 and vcc are constant values, the third item in the right-handside of the equation (3) is also a constant value; thus, even whenlowpass-filter processing is applied to the third item in the right-handside, the value does not change.

Taking the above into account, in Embodiment 1 of the present invention,by making the gain calculator 25 multiply the phase width signal D1 bythe gain Koff and then output the result of the multiplication as thecorrected signal voc corresponding to the differential detection signalerror voe, and by subtracting the corrected signal voc from the signalvof obtained by applying lowpass-filter processing to the differentialdetection signal vo, the common mode rejection is performed.

The second electric potential v2 approximately coincides with themultiplication product of the first electric potential v1 and the phasewidth signal D1. In the case where the first electric potential v1 is aconstant value, the gain Koff is also a constant value; however, whenthe first electric potential v1 changes, the differential detectionsignal error voe also changes in accordance with the change in the firstelectric potential v1. In such a case as the first electric potential v1changes, the gain Koff may change in accordance with the first electricpotential v1.

As described above, in the electric-power conversion apparatus accordingto Embodiment 1 of the present invention, the differential detectionsignal vo is corrected by the control signal for the chopper circuit;therefore, the current value can be calculated in accordance with theoffset change due to an in-phase voltage, whereby current detection canaccurately be performed.

In the conventional electric-power conversion apparatus disclosed inPatent Document 1, when the upper stage power semiconductor device turnson, no current is supplied to the detection resistor, whereby in orderto obtain a current, it is required to provide a duration in which theupper stage power semiconductor device is turned off; however, inEmbodiment 1, because the current sense resistor is connected betweenthe second electric potential and the third electric potential, thecurrent supplied from the second electric potential v2 to the thirdelectric potential v3 can be obtained, regardless of whether or not theupper stage power semiconductor device or the lower stage powersemiconductor device is on.

Because the calculation means 13 calculates the corrected signal voc,based on a value proportional to the phase width signal D1, which is thecontrol signal for the chopper circuit, it is made possible to performcommon mode rejection by implementing correction in accordance with thechange in the phase width signal D1; therefore, the current suppliedfrom the second electric potential to the third electric potential canaccurately be obtained.

In the conventional electric-power conversion apparatus disclosed ineach of Patent Documents 1 and 2, in some cases, the current passingthrough the current sense resistor and the output current of the choppercircuit coincide with each other, and in some cases, they do notcoincide with each other, depending on whether or not the choppercircuit is on. Accordingly, in such a conventional electric-powerconversion apparatus as described above, in the case where a lowpassfilter is applied to the differential detection signal and the timeconstant is set to be approximately equal to the cycle of thecarrier-wave signal or larger, the value at a time when a current issupplied to the current sense resistor and the value at a time when nocurrent is supplied to the current sense resistor are mingled with eachother, and hence no current can be detected. In contrast, in Embodiment1, the current passing through the current sense resistor and the outputcurrent of the chopper circuit are the same as each other, and norestriction is imposed on the time constant of the lowpass filter thatis applied to the differential detection signal.

Moreover, in the calculation means 13, when a filter is applied to thedifferential detection signal vo, the time constant of the filter is setto a constant raging from a value that is approximately equal to thecycle of the carrier-wave signal to a value that is approximately 20times as large as the cycle; thus, because being suppressed fromfluctuating when the switching signal becomes on/off, the current valuei1 can be corrected with the phase width signal. As a result, even whenthe differential amplification signal is analogue/digital-convertedasynchronously with the carrier wave, an accurate current detectionvalue can be obtained. The value of K2 may be obtained either from theresistance value of the differential detection circuit or throughexperimental measurement of K2.

Embodiment 2

In Embodiment 1, the lower stage power semiconductor device included inthe chopper circuit is formed of a diode, and the upper stage powersemiconductor device is turned on or off based on the switching signalS1 outputted by the calculation means; however, the upper stage powersemiconductor device and the lower stage power semiconductor deviceincluded in the chopper circuit may be configured in such a way as to beturned on or off based on the switching signals G1 and G2, respectively,outputted by the calculation means.

FIG. 4 is a diagram illustrating the overall configuration of this kindof electric-power conversion apparatus according to Embodiment 2 of thepresent invention; in the drawing, the same reference characters asthose in Embodiment 1 denote the same or similar constituent elements.

In FIG. 4, an electric-power conversion apparatus 15 a is provided witha chopper circuit 16 a, the differential detection circuit 6, acalculation means 13 a, the phase width signal generator 14, and thecurrent sense resistor 4; as the control signals for the chopper circuit16 a, there exist the phase width signal D1 outputted by the phase widthsignal generator 14 and switching signals G1 and G2 outputted by thecalculation means 13 a. The phase width signal D1 corresponds to theratio of the conduction duration of an upper stage power semiconductordevice 2 a to the conduction duration of a lower stage powersemiconductor device 3 a and varies ranging from “0” to “1”; the upperstage power semiconductor device 2 a and the lower stage powersemiconductor device 3 a are included in the chopper circuit 16 a. Thephase width signal D1 is also referred to as a duty ratio. The switchingsignal G1 is a signal for turning on or off the upper stage powersemiconductor device 2 a included in the chopper circuit 16 a; theswitching signal G2 is a signal for turning on or off the lower stagepower semiconductor device 3 a included in the chopper circuit 16 a.

The negative electrode of the DC voltage source 1 having the voltagedifference v1 is connected with the ground potential; the positiveelectrode of the DC voltage source 1 is connected with one of theterminals of the upper stage power semiconductor device 2 a included inthe chopper circuit 16 a. The upper stage power semiconductor device 2 aturns on and off when the switching signal G1 is “TRUE” and “FALSE”,respectively. One of the terminals of the lower stage powersemiconductor device 3 a and one of the terminals of the current senseresistor 4 having a resistance value of R0 [Ω] are connected with theother terminal of the upper stage power semiconductor device 2 a; theother terminal of the lower stage power semiconductor device 3 a isconnected with the ground potential. The lower stage power semiconductordevice 3 a turns on and off when the switching signal G2 is “TRUE” and“FALSE”, respectively.

In Embodiment 2, the upper stage power semiconductor device 2 a and thelower stage power semiconductor device 3 a are each formed of asemiconductor switching device such as a MOS-FET or an IGBT. Even in thecase where as described above, the lower stage power semiconductordevice 3 a included in the chopper circuit 16 a is formed of a MOS-FET,the same operation as in Embodiment 1 can be performed by inputting theswitching signal G2 thereto in such a way as described later.

Comparing the case where the lower stage power semiconductor device isformed of a diode with the case where the lower stage powersemiconductor device is formed of a MOS-FET, a diode has the advantageof being cheaper than a MOS-FET; in contrast, a MOS-FET has theadvantage in that the conduction loss and the heating thereof aresmaller than those of a diode.

FIG. 5 is a diagram illustrating the internal configuration of thecalculation means 13 a according to Embodiment 2. The calculation means13 a calculates the switching signals G1 and G2 for turning on or offthe upper stage power semiconductor device 2 a and the lower stage powersemiconductor device 3 a, respectively, based on a desired phase widthsignal D1 obtained from the phase width signal generator 14, calculatesthe current i0 passing through the current sense resistor 4, based onthe differential detection signal vo, and outputs the current i0 as acalculation current i1. In the calculation means 13 a, a delay device22, an AND circuit 23, and a NOR circuit 24 are added to the calculationmeans 13 of Embodiment 1.

The delay device 22 outputs, as a signal S2, the signal obtained bydelaying the signal S1 by a predetermined time Td. The predeterminedtime Td [sec], which is known as a dead time, is a time for preventingthe upper stage power semiconductor device 2 a and the lower stage powersemiconductor device 3 a from simultaneously turning on, causing ashort-circuit which produces an excessive current between the firstelectric potential and the reference electric potential. In Embodiment2, the delay time Td is set to 5×10⁻⁶ [sec].

The AND circuit 23 calculates the logical multiplication product of thesignal S1 and the signal S2 and outputs the product, as the switchingsignal G1. The NOR circuit 24 calculates the negative OR of the signalS1 and the signal S2 and outputs the negative OR, as the switchingsignal G2.

FIG. 6 is a set of charts for explaining the calculation, by thecalculation means 13 a, of the switching signals G1 and G2 based on adesired phase width signal D1 and a carrier-wave signal C1. In FIG. 6,the top chart represents the waveforms of the phase width signal D1 andthe carrier-wave signal C1. The phase width signal D1 ranges from “0” to“1”. The carrier-wave signal C1 has the minimum value “0”, the maximumvalue “1”, and the cycle Tc. The comparator 21 compares the level of thephase width signal D1 and the level of the carrier-wave signal C1; inthe case where D1 C1, the signal S1 becomes “TRUE”, and in the casewhere D1<C1, the signal S1 becomes “FALSE”. The signal S2 is a signalobtained by delaying the signal S1 by the predetermined time Td. Theswitching signal G1 is a signal obtained by calculating the logicalmultiplication product (AND) of the signal S1 and the signal s2; theswitching signal G2 is a signal obtained by calculating the negative OR(NOR) of the signal S1 and the signal s2. By calculating the switchingsignals G1 and G2 in this way, there is obtained the relationshipbetween the switching signals G1 and G2 that are represented in thefourth top chart and the fifth top chart of FIG. 6. That is to say, theswitching signals G1 and G2 alternatively become on, and in order toprevent the switching signals G1 and G2 from simultaneously becoming on,there is provided a duration in which the switching signals G1 and G2simultaneously become off just for the delay time Td.

As described above, in the electric-power conversion apparatus accordingto Embodiment 2 of the present invention, the upper stage powersemiconductor device and the lower stage power semiconductor deviceincluded in the chopper circuit are each formed of a power semiconductorswitching device that is turned on or off based on the switching signalsG1 and G2, respectively; the same operation and effect as thosedemonstrated in Embodiment 1 can be obtained also with Embodiment 2.

Embodiment 3

In the calculation means 13 a of Embodiment 2, the calculation currenti1 is continuously updated; however, the calculation means 13 a may bereplaced by a calculation means in which the calculation current i1 isdiscretely updated at an arbitrary timing.

FIG. 7 is a diagram illustrating the internal configuration of acalculation means 13 b according to Embodiment 3 of the presentinvention; in the drawing, the same reference characters as those inEmbodiment 2 denote the same or similar constituent elements.

In FIG. 7, a holder 30 holds the phase width signal D1 and the outputvof of the lowpass filter 26 at an arbitrary timing and outputs a signalvoh. The calculation means 13 b is the same as the calculation means 13a of Embodiment 2, excluding the holder 30. Because both the phase widthsignal D1 and the output vof of the lowpass filter 26 continuouslychange, no restriction is imposed on the timing when the holder 30performs holding operation; thus, the calculation current i1 isdiscretely updated at an arbitrary timing.

In such conventional electric-power conversion apparatuses as disclosedin Patent Documents 1 and 2, in the case where the calculation currentis discretely updated, there is imposed restriction that thedifferential detection signal should be held in synchronization with thetiming when the lower stage semiconductor device in the chopper circuitturns on, because values at a time when no current is supplied to thecurrent sense resistor may be held; however, in an electric-powerconversion apparatus according to Embodiment 3, there exists no suchrestriction, and the calculation means 13 b may analogue/digital-convertthe signal vof and the phase width signal D1 asynchronously with thecarrier-wave signal C1. As a result, an accurate current detection valuecan be obtained even with an inexpensive holder whose holding timing isinaccurate.

Embodiment 4

In each of the foregoing embodiments, the calculation means obtains thecorrection amount for the calculation current i1, based on the phasewidth signal; however, the calculation means may be replaced by acalculation means that obtains the correction amount for the calculationcurrent i1, based on the switching signals G1 and G2 or the signal S1outputted by the comparator 21.

FIG. 8 is a diagram illustrating the internal configuration of acalculation means 13 c according to Embodiment 4 of the presentinvention; in the drawing, the same reference characters as those inEmbodiments 1 and 3 denote the same or similar constituent elements.

In FIG. 8, a first correction amount setter 31 outputs a firstcorrection amount voff1; a second correction amount setter outputs asecond correction amount voff2. A selector 33 outputs, as the correctedsignal voc, the first correction amount voff1 in the case where thesignal S1 is “TRUE”, and outputs the second correction amount voff2 inthe case where the signal S1 is “FALSE”.

FIG. 9 is a set of charts representing the relationship among the timeand the respective signals in the calculation means 13 c according toEmbodiment 4. In FIG. 9, the top chart represents the phase width signalD1 and the carrier-wave signal C1; the second top chart represents thesignal S1; the third top chart represents the current i0; the fourth topchart represents the differential detection signal vo; the bottom chartrepresents the differential detection signal error voe. The secondelectric potential v2 is equal to the first electric potential v1 whenthe upper stage power semiconductor device 2 is on, and is equal to thereference electric potential when the lower stage power semiconductordevice 3 is on. In other words, if the delay time Td is neglected, thesecond electric potential v2 is equal to the first electric potential v1when the signal S1 is “TRUE”, and is equal to the reference electricpotential when the signal S1 is “FALSE”. Accordingly, in the case wherethe foregoing equation (8) is satisfied with regard to the differentialdetection signal error voe, two cases, i.e., the case where the signalS1 is “TRUE” and the case where the signal S1 is “FALSE” may be takeninto consideration with regard to the differential detection signalerror voe.

In the case where the signal S1 is “TRUE”, the second electric potentialv2 is equal to the first electric potential v1; therefore, thedifferential detection signal error voe is (K2×v1), which is obtained bysubstituting v1 for v2 in the equation (8). In the case where the signalS1 is “FALSE”, the second electric potential v2 is equal to thereference electric potential; therefore, the differential detectionsignal error voe is “0” (=K2×0), which is obtained by substituting “0”for v2 in the equation (8).

Thus, the first correction amount setter 31 is set in such a way as tooutput K2×v1 and the second correction amount setter 32 is set in such away as to output “0”, so that the selector 33 can output the accuratecorrected signal voc.

The signals S1 and S2 and the switching signals G1 and G2 are in therelationship represented in FIG. 6, and, in general, the delay time Tdis negligible; thus, in the case where as the input of the selector 33,any one of the switching signals G1 and G2 and the signal S2 is utilizedinstead of the signal S1, the same effect can be demonstrated.

As the setting values for the first correction amount voff1 and thesecond correction amount voff2, the values of the offset signal voffoutputted by the offset setter 28 is incorporated, so that the offsetsetter 28 can be omitted. The first correction amount voff1 is acorrection amount at a time when the second electric potential v2 isequal to the first electric potential v1. In the case where the firstelectric potential v1 is a constant value, the first correction amountvoff1 may be set to be constant; however, it goes without saying that inthe case where the first electric potential v1 changes, the firstcorrection amount voff1 may be made to change in accordance with thefirst electric potential v1.

As described above, in the calculation means 13 c according toEmbodiment 4 of the present invention, one of the first correctionamount and the second correction amount is selected in accordance withwhether or not the switching signals G1 and G2 turns on or off;therefore, there is demonstrated an effect that the calculation of acorrection amount, which is implemented every phase width signal D1, isomitted, and accurate current calculation can be performed with only twocorrection amounts.

Embodiment 5

In the calculation means 13 c of Embodiment 4, the calculation currenti1 is continuously updated; however, the calculation means 13 c may bereplaced by a calculation means in which the calculation current i1 isdiscretely updated.

FIG. 10 is a diagram illustrating the internal configuration of acalculation means 13 d according to Embodiment 5 of the presentinvention; in the drawing, the same reference characters as those inEmbodiment 4 denote the same or similar constituent elements.

In FIG. 10, a selector 33 d holds the differential detection signal voat a timing described later, based on the phase width signal D1 and thecarrier-wave signal C1, and outputs the differential detection signalvo, as the signal voh; concurrently, the selector 33 d selects one ofthe first correction amount voff1 and the second correction amount voff2and outputs it, as the corrected signal voc.

FIG. 11 is a flowchart representing the operation of the selector 33 d.In the step S100, the selector 33 d starts its operation. In the stepS101, it is determined whether or not the phase width signal D1 islarger than 0.5. In the case where it is determined that the phase widthsignal D1 is larger than 0.5, the selector 33 d waits for ΔT seconds inthe step S102. In Embodiment 5, ΔT [sec] is set in such a way that thevalue that is integer-fold as large as ΔT is equal to the half cycleTc/2 [sec] of the carrier-wave signal C1. In addition, it is desirablethat ΔT [sec] is set to be integer-fold as large as the half cycle Tc/2[sec] of the carrier-wave signal C1; however, ΔT may not be integer-foldas large as the half cycle Tc/2 [sec] of the carrier-wave signal C1, aslong as ΔT is the same as or smaller than Tc/2. In the step S103, it isdetermined whether or not the carrier-wave signal C1 is “0”. In the casewhere the carrier-wave signal C1 is “0”, the first correction amountvoff1 is selected, as the corrected signal voc, in the step S104; then,in the step S105, the signal voh is updated and becomes the signal vo ata time point when the carrier-wave signal C1 is “0”. In the case whereit is determined, in the step S103, that the carrier-wave signal C1 isnot “0”, the process of the step S102 is implemented. In the case whereit is determined, in the step S101, that the phase width signal D1 isthe same as or smaller than 0.5, the selector 33 d waits for ΔT secondsin the step S106.

In the step S107, it is determined whether or not the carrier-wavesignal C1 is “1”. In the case where the carrier-wave signal C1 is “1”,the second correction amount voff2 is selected, as the corrected signalvoc, in the step S108; then, in the step S105, the signal voh is updatedand becomes the signal vo at a time point when the carrier-wave signalC1 is “1”. In the case where it is determined, in the step S107, thatthe carrier-wave signal C1 is not “1”, the process of the step S106 isimplemented.

According to the flowchart in FIG. 11, in the case where the phase widthsignal D1 is larger than 0.5, the differential detection signal is heldat a timing when the carrier-wave signal C1 becomes “0” (when thetriangular-shaped carrier-wave signal C1 is an extreme bottom); in thecase where the phase width signal D1 is the same as or smaller than 0.5,the differential detection signal is held at a timing when thecarrier-wave signal C1 becomes “1” (when the triangular-shapedcarrier-wave signal C1 is a peak).

FIG. 12 is a set of charts representing the relationship among the timeand the respective signals in Embodiment 5. In FIG. 12, the top chartrepresents the phase width signal D1 and the carrier-wave signal C1; thesecond top chart represents the signal S1; the third top chartrepresents the current i0; the fourth top chart represents thedifferential detection signal vo and the held signal voh; the bottomchart represents the corrected signal voc. In a duration, the phasewidth signal D1 is the same as or smaller than 0.5; in a duration B, thephase width signal D1 is larger than 0.5.

In the duration A where the phase width signal D1 is the same as orsmaller than 0.5, the differential detection signal vo is held at atiming when the carrier-wave signal C1 becomes “1”, i.e., when thetriangular-shaped carrier-wave signal C1 is a peak, and then isoutputted as the signal voh. Then, in the duration A, the secondcorrection amount voff2 is selected and is outputted as the correctedsignal voc.

In the duration B where the phase width signal D1 is larger than 0.5,the differential detection signal vo is held at a timing when thecarrier-wave signal C1 becomes “0”, i.e., when the triangular-shapedcarrier-wave signal C1 is an extreme bottom, and then is outputted asthe signal voh. Then, in the duration B, the first correction amountvoff1 is selected and is outputted as the corrected signal voc.

In the duration A where the phase width signal D1 is the same as orsmaller than 0.5, the FALSE duration of the signal S1 is longer than theTRUE duration thereof; at the timing when the carrier-wave signal C1 is“1”, the signal S1 is FALSE. In the duration B where the phase widthsignal D1 is larger than 0.5, the FALSE duration of the signal S1 isshorter than the TRUE duration thereof; at the timing when thecarrier-wave signal C1 is “0”, the signal S1 is TRUE. With suchoperation as described above, the differential detection signal vo isheld while the signal S1 is TRUE in the duration B where the TRUE pulseduration of the signal S1 is longer than the FALSE pulse durationthereof or while the signal S1 is FALSE in the duration A where the TRUEpulse duration of the signal S1 is shorter than the FALSE pulse durationthereof, so that the signal voh is obtained.

With such a configuration as described above, in the calculation means13 d according to Embodiment 5, in the case where the phase width signalD1 is larger than 0.5, the differential detection signal vo is heldduring the duration where the upper stage power semiconductor device 2 ais on, i.e., where the signal S1 is TRUE. In the case where the phasewidth signal D1 is the same as or smaller than 0.5, the differentialdetection signal vo is held during the duration where the lower stagepower semiconductor device is on, i.e., where the signal S1 is FALSE. Asa result, the differential detection signal vo is held always during theduration where the signal S1 maintains TRUE longer than FALSE or wherethe signal S1 maintains FALSE longer than TRUE. In other words, it isnot required to hold the differential detection signal vo when TRUE andFALSE alternate with each other in a short time; therefore, even whenthe holding timing is slightly displaced from the peak or the extremebottom of the triangular-shaped carrier-wave signal C1, the state of thesignal S1 does not change.

In the conventional electric-power conversion apparatus in each offoregoing Patent Documents 1 and 2, it is required to hold thedifferential detection signal at a timing when the lower stage powersemiconductor device becomes on; however, when the phase width signal D1approaches to “1”, the duration where the lower stage powersemiconductor device is on is short; thus, there has been a problem thatthere is required a high-accurate holding timing for the differentialdetection signal. In the electric-power conversion apparatus accordingto Embodiment 5, it is not required to hold the differential detectionsignal vo when TRUE and FALSE alternate with each other in a short time;therefore, because the tolerance range of the holding timing for thedifferential detection signal vo becomes wide, a current value can beobtained even with an inexpensive circuit that produces a delay timecaused through processing.

In Embodiment 5, as an example, there has been described a case where inthe step S101, the branch condition for the phase width signal D1 is setto 0.5; however, it goes without saying that the branch condition forthe phase width signal D1 may be a numerical value other than 0.5, aslong as it is in the range between “0” and “1”.

Embodiment 6

In the calculation means 13 d in Embodiment 5, by holding thedifferential detection signal vo, the calculation current i1 isoutputted; however, the calculation means 13 d may be replaced by acalculation means in which instead of the differential detection signalvo, the signal vof obtained by applying a lowpass filter to thedifferential detection signal vo is held.

FIG. 13 is a diagram illustrating the internal configuration of acalculation means 13 e according to Embodiment 6 of the presentinvention; in the drawing, the same reference characters as those inEmbodiment 5 denote the same or similar constituent elements.

In FIG. 13, a lowpass filter 40 applies filtering with a time constantdescribed later to the differential detection signal vo and inputs theresult, as the signal vof, to a selector 33 e. The selector 33 e holdsthe signal vof at a timing described later, based on the phase widthsignal D1 and the carrier-wave signal C1, and outputs the signal vof, asthe signal voh; concurrently, the selector 33 e selects one of the firstcorrection amount voff1 and the second correction amount voff2 andoutputs it, as the corrected signal voc.

FIG. 14 is a flowchart representing the operation of the selector 33 e;in the drawing, the same reference characters as those in Embodiment 5denote the same or similar constituent elements.

After the processing of the step S104 or the step S108, the selector 33e waits for ΔT2 [sec] in the step S110. Through the step S110, thesignal voh from the selector 33 e is updated at a timing that is ΔT2behind the timing of the peak or the extreme bottom of thetriangular-shaped carrier-wave signal C1. ΔT2 is set to be a valueranging from “0” to a quarter of the cycle of the carrier-wave signalC1, i.e., Tc/4 [sec]. In particular, in the case where ΔT2 is 0 [sec],the selector 33 e becomes the same as the selector 33 d.

FIG. 15 is a set of charts representing the relationship among the timeand the respective signals in Embodiment 6. In FIG. 15, the top chartrepresents the phase width signal D1 and the carrier-wave signal C1; thesecond top chart represents the signal S1; the third top chartrepresents the current i0; the fourth top chart represents thedifferential detection signal error voe, the fifth top chart representsthe signal voef obtained by filtering the differential detection signalerror voe with a filter having a time constant the same as that of thelowpass filter 40, and the bottom chart represents the differentialdetection signal vo and the signal voh obtained through holding. Asrepresented in the third top chart of FIG. 15, an oscillation, which iscalled ringing, occurs in the current i0 at a timing when the signal S1changes, i.e., at a timing when the power semiconductor device turns onor off. If the selector holds a signal at a timing when the ringingexists, the ringing causes an error in the calculation current i1outputted by the calculation means. The lowpass filter 40 in Embodiment6 operates in such a way as to eliminate the effect of the ringing.

Here, the time constant of the lowpass filter 40 will be considered. Thefourth top chart of FIG. 15 represents the differential detection signalerror voe; the fifth top chart represents the waveform of the signalvoef obtained by filtering the differential detection signal error voewith a filter having a time constant the same as that of the lowpassfilter 40. As represented in FIG. 15, in the case where the phase widthsignal D1 is larger than 0.5, the selector 33 e holds the signal vof, asdescribed above, at a timing ΔT2 [sec] behind the timing when thetriangular-shaped carrier-wave signal C1 is an extreme bottom. ΔT2 isset to be a value ranging from 0 [sec] to a quarter of the cycle of thecarrier-wave signal C1, i.e., Tc/4 [sec]; therefore, the selector 33 eholds the signal vof at a given timing within the duration C in FIG. 15.In other words, by changing ΔT2 within a range from “0” to Tc/4 [sec],the signal vof is held at an arbitrary timing within the duration C inthe FIG. 15; in the example represented in FIG. 15, by setting ΔT2 toTc/8 [sec], the signal vof is held at the timing indicated by the mark“▴”.

In the case where the phase width signal D1 is larger than 0.5 and theerror voef included in the signal vof held by the selector 33 e is avalue corresponding to the first correction amount voff1, the firstcorrection amount voff1 is an appropriate value, as the correctionamount selected by the selector 33 e. As can be seen from the fifth topchart of FIG. 15, in the case where the phase width signal D1 is largerthan 0.5, the signal vof is held during a duration in which the signalS1 is TRUE; thus, the signal voef takes a value existing in a durationbetween 0 [sec] and Tc/2 [sec] from a timing when the signal S1 changesfrom FALSE to TRUE. Therefore, the time constant of the lowpass filter40 may be set to be the same as or smaller than half of the cycle of thecarrier-wave signal C1. In addition, by setting the time constant of thelowpass filter 40 to be the same as or larger than 1/20 times as largeas the cycle of the carrier-wave signal C1, the change of vo with regardto the carrier-wave signal C1 can sufficiently be recognized. That is tosay, the time constant of the lowpass filter 40 may be set to be between1/20 times as large as the cycle of the carrier-wave signal C1 and halfof the cycle of the carrier-wave signal C1.

In particular, by holding the signal vof during the duration C, thesignal voef takes a value existing in a duration between Tc/4 [sec] andTc/2 [sec] from a timing when the signal S1 changes from FALSE to TRUE.In other words, because the duration in which the selector 33 e isallowed to hold the signal vof is the same as or longer than half of thecycle of the carrier-wave signal, even when there is provided a filterhaving a time constant between a quarter and half of the cycle of thecarrier-wave signal C1, the differential detection signal error voe isin a steady state at a holding timing; thus, it is desirable. As aresult, it is made possible to suppress detection noise caused by aringing or the like; therefore, the first correction amount voff1 is anappropriate value, as the correction amount selected by the selector 33e.

Such a conventional electric-power conversion apparatus as disclosed inPatent Documents 1 and 2 is configured in such a way that thedifferential signal is held at the timing when the lower stage powersemiconductor device becomes on; therefore, in such a case asrepresented in FIG. 15, a current is supplied to the current senseresistor only the duration when the signal S1 is FALSE. Thus, theholding duration is equal to the duration when the signal S1 is FALSE.As the phase width signal D1 approaches more to “1”, the duration ofFALSE becomes shorter. As the range of the phase width signal D1 issituated higher, the utilization rate of the electric-power conversionapparatus becomes higher; For the ordinary application, it is desired toensure the utilization rate of 0.95 or higher. In this case, theduration in which the signal S1 is FALSE becomes 1/20 times as long asthe cycle of the carrier-wave signal. Accordingly, in the case wherethere is provided a filter having a time constant between 1/20 of thecycle of the carrier-wave signal and half of the cycle of thecarrier-wave signal, the differential detection signal error voe is heldwhen it is in the transient state; therefore, as the correction amountselected by the selector 33 e, two kinds of correction amounts, forexample, the first correction amount voff1 and the second correctionamount voff2 cannot be given. As a result, there cannot be demonstratedthe effect of suppressing noise caused by a ringing or the like.

However, in the electric-power conversion apparatus according toEmbodiment 6, the calculation current i1 is calculated based on thevalue vof obtained by applying a filter having a time constant betweenapproximately 1/20 of the cycle of the carrier-wave signal C1 andapproximately half of the cycle of the carrier-wave signal C1 to thedifferential detection signal vo; therefore, without undergoing theeffect of noise caused by a ringing or the like, the stable and accuratecalculation current i1 can be calculated.

Embodiment 7

In Embodiment 6, the calculation means is provided with the firstcorrection amount and the second correction amount, selects, as thecorrection amount for calculating the calculation current i1, one of thefirst correction amount and the second correction amount in accordancewith on/off of the switching signals G1 and G2, and performs additionand correction; however, in the case where the voltage drop generated inthe power semiconductor device is approximately in proportion to thecurrent, the electric potential v2, i.e., the differential detectionsignal error voe changes in accordance with the current that passesthrough the power semiconductor device. In the case where the voltagedrop generated in the power semiconductor device is approximately inproportion to the current, it is reflected in such a way that theamplification factor of the differential detection signal vo to thecurrent i0 changes. In the case where as the upper stage powersemiconductor device and the lower stage power semiconductor device,different types of power semiconductor devices are utilized, thecoefficient of the proportion relationship between the voltage dropgenerated in the power semiconductor device and the current i0 differsdepending on the on/off state of the switching signals G1 and G2.Accordingly, it may be allowed that one of the third correction amountand the fourth correction amount is selected in accordance with thesignal S1 that changes in synchronization with on/off of the switchingsignals G1 and G2 and then multiplication and correction are performed.

FIG. 16 is a diagram illustrating the internal configuration of acalculation means 13 f according to Embodiment 7 of the presentinvention; in the drawing, the same reference characters as those inEmbodiment 6 denote the same or similar constituent elements.

In the calculation means 13 e of Embodiment 6, the gain calculator 29multiplies the output of the adder-subtractor 27 by “K”. The calculationmeans 13 f of Embodiment 7 is provided with a third correction amountsetter 50 that outputs a third correction amount K1 and a fourthcorrection amount setter 51 that outputs a fourth correction amount K2.

When selecting the first correction amount voff1, as the correctionamount to be outputted and inputted to the adder-subtractor 27, aselector 33 f selects the third correction amount K1, as the correctionamount to be outputted and inputted to a multiplier 52; when selectingthe second correction amount voff2, as the correction amount to beoutputted and inputted to the adder-subtractor 27, the selector 33 fselects the fourth correction amount K2, as the correction amount to beoutputted and inputted to the multiplier 52.

As described above, as the correction amount for calculating thecalculation current i1, the third correction amount K1 and the fourthcorrection amount K2 for correcting the calculation current i1 in amultiplication manner are added to voff1 and voff2 for correcting thecalculation current i1 in an addition manner, so that the calculationcurrent i1 can more accurately be calculated.

Embodiment 8

In the foregoing embodiments, there has been described the differentialdetection circuit 6 in which a single operational amplifier plays theroles of differential detection and signal amplification; however, thedifferential detection circuit 6 may be replaced by a differentialdetection circuit 6 g in which there are provided two operationalamplifiers that play the roles of differential detection and signalamplification.

FIG. 17 is a diagram illustrating the internal configuration of thedifferential detection circuit 6 g according to Embodiment 8 of thepresent invention; in the drawing, the same reference characters asthose in the foregoing embodiments denote the same or similarconstituent elements.

In FIG. 17, the differential detection circuit 6 g is provided withtotally two operational amplifiers, i.e., the operational amplifier 7that plays the role of differential detection and an operationalamplifier 66 that plays the role of signal amplification. The resistors8 through 12 are selected in such a way that the following equations (9)and (10) are satisfied among the resistance values R1 through R5.

R1=R2=R3  (9)

2×R1=R4=R5  (10)

By giving the resistance values R1 through R5 of the resistors 8 through12 in this manner, the operational amplifier 7 does not amplify theelectric potential difference (v3−v2) and hence the differentialdetection can be performed. The resistance values are only two kinds ofvalues; as the resistors 11 and 12, resistors having a resistance valueof R1 are given in such a way that they are connected in series witheach other, or as the resistors 8 through 12, resistors having aresistance value of R4 are given in such a way that they are connectedin parallel with one another, so that the resistors 8 through 12 can beformed of the same resistors in the same lot. As a result, there can bereduced a detection error produced through in-phase voltages caused byunevenness in the resistance value. By giving the resistance values R10through R15 of resistors 60 through 65 in an appropriate manner, theoperational amplifier 66 can output a value obtained by amplifying theoutput of the operational amplifier 7 with a desired amplificationfactor.

As described above, the differential detection circuit according toEmbodiment 8 is configured in such a way as to be provided with totallytwo operational amplifiers, i.e., an operational amplifier that playsthe role of differential detection and an operational amplifier thatplays the role of signal amplification, the resistors 8 through 12connected with the operational amplifier that plays the role ofdifferential detection can be formed in the same lot. Because theunevenness in the resistance values of resistors in the same lot issmaller than that in the resistance values of resistors in differentlots, the unevenness in the resistance values of the resistors 8 through12 and the error caused by in-phase voltages can be reduced. As aresult, there can be demonstrated an effect that the error in thedifferential detection signal vo can be reduced.

Embodiment 9

As described above, when being mounted in a vehicle, an electric-powerconversion apparatus needs to operate at a severe temperature, at asevere humidity, in a vibration, and in a dusty condition; thus, acurrent detecting resistor is superior to a hole device as the holecurrent sensor in terms of robustness. Accordingly, in Embodiment 1, theelectric-power conversion apparatus 15 may be configured in such a waythat the first electric potential v1 is connected with the positiveelectrode of the battery 1 mounted in a vehicle, the third electricpotential v3 is connected with the electric load 5 mounted in thevehicle, and the reference electric potential GND is earthed to thevehicle body or connected with the negative electrode of the battery 1.

By being configured in this way, the electric-power conversion apparatus15 can satisfy the limiting conditions with regard to temperature,humidity, vibration, and dust, which should be cleared when mounted in avehicle, can apply a voltage lower than the battery voltage to theelectric load, and can demonstrate an effect of accurately obtaining thevalue of the current i0 to be supplied to the electric load.

Embodiment 10

In Embodiment 9, there has been dealt with a case where the firstelectric potential v1 of the electric-power conversion apparatus isconnected with the battery and the third electric potential v3 isconnected with the electric load; however, the electric-power conversionapparatus may be configured in such a way that the first electricpotential v1 is connected with the electric load, the third electricpotential v3 is connected with the positive electrode of the battery byway of a coil, and the reference electric potential is earthed to thevehicle body or connected with the negative electrode of the battery.

FIG. 18 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 10 of thepresent invention; in the drawing, the same reference characters asthose in the foregoing embodiments denote the same or similarconstituent elements.

In FIG. 18, an electric-power conversion apparatus 15 h is provided witha chopper circuit 16 h, the current sense resistor 4, the differentialdetection circuit 6, the calculation means 13, and the phase widthsignal generator 14; as the control signals for the chopper circuit 16h, there exist the phase width signal D1 outputted by the phase widthsignal generator 14 and the switching signal S1 outputted by thecalculation means 13.

The phase width signal D1 corresponds to the ratio of the conductionduration of the lower stage power semiconductor device 3 h included inthe chopper circuit 16 h to the non-conduction duration thereof andvaries ranging from “0” to “1”. A second electric potential v2 becomesequal to the reference electric potential GND during a duration in whichthe lower stage power semiconductor device 3 h is on. An electric load 5h is a load consisting of a capacitor and a resistor that are connectedin parallel with each other.

One of the terminals of the current sense resistor 4 is connected withthe connection point between the upper stage power semiconductor device2 h and the lower stage power semiconductor device 3 h that are includedin the chopper circuit 16 h; the positive electrode of a DC voltagesource 1 h is connected with the other terminal of the current senseresistor 4 by way of a coil 80. The first electric potential v1 is theelectric potential of the connection point between the chopper circuit16 h and the electric load 5 h; the second electric potential v2 is theelectric potential of the connection point between the chopper circuit16 h and the current sense resistor 4; the third electric potential v3is the electric potential of the connection point between the currentsense resistor 4 and the coil 80.

The current i0 that passes through the coil 80 always variescontinuously; therefore, when the state of the lower stage powersemiconductor device 3 h changes from “on” to “off”, the current i0 issupplied from the second electric potential v2 to the electric load 5 hby way of the upper stage power semiconductor device 2 h, which is adiode. In Embodiment 10, the upper stage power semiconductor deviceincluded in the chopper circuit 16 h is formed of a diode, which is asemiconductor rectifier; however, even when the upper stage powersemiconductor device is formed of a semiconductor switching device suchas a MOS-FET or an IGBT, operation the same as that of Embodiment 10 canbe performed, as long as appropriate switching is implemented. Comparingthe case where the upper stage power semiconductor device is formed of aMOS-FET with the case where the upper stage power semiconductor deviceis formed of a diode, a MOS-FET has the advantage in that the conductionloss and the heating thereof are smaller than those of a diode; incontrast, a diode has the advantage of being cheaper than a MOS-FET.

The switching signal S1 is a signal for turning on or off the lowerstage power semiconductor device 3 h included in the chopper circuit 16h.

With such a configuration as in Embodiment 10, the first electricpotential v1 with which the electric load 5 h is connected can be madehigher than the electric potential of the positive electrode of the DCvoltage source 1 h. In addition, when being mounted in a vehicle, anelectric-power conversion apparatus needs to operate at a severetemperature, at a severe humidity, in a vibration, and in a dustycondition; thus, a current detecting resistor is superior to a holedevice as the hole current sensor in terms of robustness. Accordingly,the first electric potential v1 of the electric-power conversionapparatus 15 h is connected with the electric load 5 h mounted in avehicle, and by way of the coil 80, the third electric potential v3 isconnected with the positive electrode of a battery mounted, as the DCvoltage source 1 h, in the vehicle, so that the electric-powerconversion apparatus can satisfy the limiting conditions with regard totemperature, humidity, vibration, and dust, which should be cleared whenmounted in a vehicle, can apply a voltage higher than the electricpotential of the positive electrode of the DC voltage source 1 h to theelectric load 5 h, and can demonstrate an effect of accurately obtainingthe value of the current i0 to be supplied from the DC voltage source 1h.

Embodiment 11

In each of the foregoing embodiments, there has been dealt with a casewhere the electric load is formed of a resistor, a coil, or a capacitor;however, the electric load may also be a DC rotating machine.

FIG. 19 is a diagram illustrating the overall configuration of anelectric-power conversion apparatus according to Embodiment 11 of thepresent invention; in the drawing, the same reference characters asthose in the foregoing embodiments denote the same or similarconstituent elements.

In FIG. 19, the DC voltage source 1, a DC rotating machine 70, and avoltage command generator 71 are connected with an electric-powerconversion apparatus 15 i. The electric-power conversion apparatus 15 iis configured with a first chopper circuit 16 x, a second choppercircuit 16 y, a phase width calculator 72, the resistor 4, thedifferential detection circuit 6, a first calculation means 13 x, and asecond calculation means 13 y.

The DC rotating machine 70 has two terminals for the x-phase and they-phase; the x-phase is connected with the chopper circuit 16 x by wayof the current sense resistor 4. The y-phase of the DC rotating machine70 is connected with the chopper circuit 16 y. The voltage commandgenerator 71 outputs a command value vx* for a voltage to be applied bythe chopper circuit 16 x to the x-phase of the DC rotating machine 70and a command value vy* for a voltage to be applied by the choppercircuit 16 y to the y-phase of the DC rotating machine 70.

The command values vx* and each vary from −0.5×v1 [V] to +0.5×v1 [V].The phase width calculator 72 preliminarily stores the electricpotential difference v1 between the terminals of the DC voltage source 1and calculates phase width signals D1 x and D1 y, based on the followingequations (11) and (12).

D1x=vx*÷v1+0.5  (11)

D1y=vy*÷v1+0.5  (12)

As can be seen from the range of the command values vx* and vy*, theranges of the phase width signals D1 x and D1 y are each from “0” to“1”. The calculation means 13 x is configured in the same manner as thecalculation means 13 described in each of the foregoing embodiments.Based on the phase width signal D1 x, the calculation means 13 x outputsa switching signal G1 x for turning on or off an upper stage powersemiconductor device 2 x in the first chopper circuit 16 x and aswitching signal G2 x for turning on or off a lower stage powersemiconductor device 3 x, and calculates the calculation current i1,based on the differential detection signal vo obtained from thedifferential detection circuit 6.

The calculation means 13 y is a calculation means in which thecalculation of the calculation current i1 performed in the calculationmeans 13 x is removed; based on the phase width signal D1 y, thecalculation means 13 y outputs a switching signal Gly for turning on oroff an upper stage power semiconductor device 2 y in the second choppercircuit 16 y and a switching signal G2 y for turning on or off a lowerstage power semiconductor device 3 y.

The connection among the chopper circuit 16 x, the current senseresistor 4, the differential detection circuit 6, and the calculationmeans 13 x is the same as the connection among the chopper circuit 16,the current sense resistor 4, the differential detection circuit 6, andthe calculation means 13; therefore, the chopper circuits in Embodiment11 can operate in the same principle as the chopper circuit inEmbodiment 1 operates. Accordingly, it goes without saying that even inthe case where the configuration according to each of Embodiments 2through 10 is utilized, it is made possible to obtain the calculationcurrent i1 while driving the DC rotating machine 70.

In addition, when being mounted in a vehicle, an electric-powerconversion apparatus needs to operate at a severe temperature, at asevere humidity, in a vibration, and in a dusty condition; thus, acurrent detecting resistor is superior to a hole device as the holecurrent sensor in terms of robustness. Accordingly, the electric-powerconversion apparatus 15 i may be configured in such a way that as the DCvoltage source 1, there is provided a battery mounted in a vehicle, thefirst electric potential v1 of the electric-power conversion apparatus15 i is connected with the positive electrode of the battery, the thirdelectric potential v3 of the electric-power conversion apparatus 15 i isconnected with the DC rotating machine mounted in the vehicle, and thereference electric potential GND is earthed to the vehicle body orconnected with the negative electrode of the battery.

By making the connection in this way, it is made possible to obtain anelectric-power conversion apparatus that satisfies the limitingconditions with regard to temperature, humidity, vibration, and dust,which should be cleared when mounted in a vehicle, applies a voltagelower than the battery voltage to the DC rotating machine 70, andaccurately obtains the value of a current to be supplied to the DCrotating machine 70.

Embodiment 12

In Embodiment 11, there has been dealt with a case where as the electricload, a DC rotating machine is connected; however, the electric load mayalso be an AC rotating machine.

FIG. 20 is a diagram illustrating the overall configuration of this kindof electric-power conversion apparatus according to Embodiment 12 of thepresent invention; in the drawing, the same reference characters asthose in the foregoing embodiments denote the same or similarconstituent elements.

In FIG. 20, the DC voltage source 1, an AC rotating machine 70 j, and avoltage command generator 71 j are connected with an electric-powerconversion apparatus 15 j. The electric-power conversion apparatus 15 jis configured with a first chopper circuit 16 u, a second choppercircuit 16 v, a third chopper circuit 16 w, a first calculation means 13u, a second calculation means 13 v, a third calculation means 13 w, aphase width calculator 72 j, a first current sense resistor 4 u, asecond current sense resistor 4 v, a first differential detectioncircuit 6 u and a second differential detection circuit 6 v.

The connection among the first chopper circuit 16 u, the first currentsense resistor 4 u, the first differential detection circuit 6 u, andthe first calculation means 13 u is the same as the connection among thechopper circuit 16, the current sense resistor 4, the differentialdetection circuit 6, and the calculation means 13; therefore, thechopper circuit in Embodiment 12 can operate in the same principle asthe chopper circuit in Embodiment 1 operates. The same applies to theconnection among the second chopper circuit 16 v, the second currentsense resistor 4 v, the second differential detection circuit 6 v, andthe second calculation means 13 v. In addition, the connection betweenthe third chopper circuit 16 w and the third calculation means 13 w isthe same as the connection between the second chopper circuit 16 y andthe second calculation means 13 y in Embodiment 11; therefore, thechopper circuits in Embodiment 12 can operate in the same principle asthe chopper circuits in Embodiment 11 operate. Moreover, it goes withoutsaying that when the configuration according to each of Embodiments 2through 11 is utilized to drive the AC rotating machine 70 j, it is madepossible to obtain the high-accuracy calculation current i1.

In addition, when being mounted in a vehicle, an electric-powerconversion apparatus needs to operate at a severe temperature, at asevere humidity, in a vibration, and in a dusty condition; thus, acurrent detecting resistor is superior to a hole device as the holecurrent sensor in terms of robustness. Accordingly, the electric-powerconversion apparatus 15 j may be configured in such a way that as the DCvoltage source 1, there is provided a battery mounted in a vehicle, thefirst electric potential v1 of the electric-power conversion apparatus15 j is connected with the positive electrode of the battery, the thirdelectric potential v3 of the electric-power conversion apparatus 15 j isconnected with the AC rotating machine mounted in the vehicle, and thereference electric potential GND is earthed to the vehicle body orconnected with the negative electrode of the battery. By making theconnection in this way, it is made possible to obtain an electric-powerconversion apparatus that satisfies the limiting conditions with regardto temperature, humidity, vibration, and dust, which should be clearedwhen mounted in a vehicle, applies a voltage lower than the batteryvoltage to the AC rotating machine 70 j, and accurately obtains thevalue of a current to be supplied to the AC rotating machine 70 j. Inaddition, in Embodiment 12, there has been dealt with a case where thereare detected currents of two phases among the currents of three phasesof the AC rotating machine 70 j; however, it goes without saying that inorder to detect the respective currents of the three phases, therespective detection resistors and differential detection circuits ofthe three phases may be provided.

What is claimed is:
 1. An electric-power conversion apparatuscomprising: a chopper circuit; a current sense resistor that detects theoutput current of the chopper circuit; a differential detection circuitthat outputs, as a differential detection signal, the electric potentialdifference across the current sense resistor; and a calculation meansthat corrects the differential detection signal from the differentialdetection circuit by use of a control signal for the chopper circuit soas to calculate the output current of the chopper circuit.
 2. Theelectric-power conversion apparatus according to claim 1, wherein thecalculation means calculates a switching signal for turning on or offthe chopper circuit based on a phase width signal, which is a controlsignal for the chopper circuit, and calculates the output current of thechopper circuit based on the differential detection signal; wherein thechopper circuit is configured with an upper stage power semiconductordevice connected between a first electric potential and a secondelectric potential and a lower stage power semiconductor deviceconnected between the second electric potential and a reference electricpotential; wherein the current sense resistor is connected between thesecond electric potential and the third electric potential; and whereinthe differential detection circuit detects the electric potentialdifference between the second electric potential and the third electricpotential.
 3. The electric-power conversion apparatus according to claim2, wherein the calculation means is provided with a first correctionamount and a second correction amount and selects, as a correctionamount for calculating the output current of the chopper circuit, one ofthe first correction amount and the second correction amount inaccordance with on/off of the switching signal.
 4. The electric-powerconversion apparatus according to claim 3, wherein the calculation meansis provided with a carrier-wave generator that generates a carrier wavefor obtaining the switching signal and a holder that holds thedifferential detection signal in synchronization with the carrier wave,and selects one of the first correction amount and the second correctionamount in accordance with the switching signal at a time point when thedifferential detection signal is held.
 5. The electric-power conversionapparatus according to claim 4, wherein in the case where the phasewidth signal is larger than a predetermined value, the calculation meansholds the differential detection signal during a duration in which theupper stage power semiconductor device is on and calculates the outputcurrent based on the first correction amount, and in the case where thephase width signal is smaller than the predetermined value, thecalculation means holds the differential detection signal during aduration in which the lower stage power semiconductor device is on andcalculates the output current based on the second correction amount. 6.The electric-power conversion apparatus according to claim 5, whereinthe calculation means calculates the output current, based on a value ofthe differential detection signal obtained by applying a filter having atime constant between approximately 1/20 of the cycle of the carrierwave and approximately half of the cycle of the carrier wave to thedifferential detection signal.
 7. The electric-power conversionapparatus according to claim 3, wherein the calculation means isprovided with a third correction amount and a fourth correction amount;wherein in the case where, as the correction amount for calculating theoutput current of the chopper circuit, the first correction amount isselected, the calculation means selects the third correction amount andperforms multiplication; and wherein in the case where, as thecorrection amount for calculating the output current of the choppercircuit, the second correction amount is selected, the calculation meansselects the fourth correction amount and performs multiplication.
 8. Theelectric-power conversion apparatus according to claim 1, wherein thecalculation means calculates a correction amount for the differentialdetection signal based on a value proportional to the phase widthsignal, which is a control signal for the chopper circuit.
 9. Theelectric-power conversion apparatus according to claim 8, wherein thecalculation means is provided with a carrier-wave generator thatgenerates a carrier wave for obtaining a switching signal that turns onor off the chopper circuit, and calculates the output current, based ona value of the differential detection signal obtained by applying afilter having a time constant between approximately 1/20 of the cycle ofthe carrier wave and approximately half of the cycle of the carrier waveto the differential detection signal.
 10. The electric-power conversionapparatus according to claim 2, wherein the first electric potential ofthe electric-power conversion apparatus is connected with the positiveelectrode of a battery mounted in a vehicle, the third electricpotential is connected with an electric load mounted in the vehicle, andthe reference electric potential is earthed to the vehicle body orconnected with the negative electrode of the battery.
 11. Theelectric-power conversion apparatus according to claim 2, wherein thefirst electric potential of the electric-power conversion apparatus isconnected with an electric load, the third electric potential isconnected with the positive electrode of a battery mounted in a vehicle,and the reference electric potential is earthed to the vehicle body orconnected with the negative electrode of the battery.
 12. Theelectric-power conversion apparatus according to claim 2, wherein thedifferential detection circuit is formed of a circuit that isnon-insulated from the reference electric potential.